3

The procedure is always the same. You need to compute the expectation $E\{|A_k|^2\}$, where $A_k$ are the complex symbols of the constellation: $$E\{|A_k|^2\}=\sum_kP_k|A_k|^2\tag{1}$$ $P_k$ is the probability that the $k^{th}$ symbol is chosen. Usually you can assume that all symbols are equally likely, i.e., $P_k=1/M$, where $M$ is the number of symbols.


3

Got me at that one! The "OFDM symbol acquisition" block is in fact not from gr-digital (where your other OFDM blocks come frome), but from gr-dtv, where it is used to capture DVB-T signals, if I remember correctly. It might be very DVB-specific! Let us have a look at the dvbt_rx_8k.grc example from gr-dtv (or, at least, the top half): So your understanding ...


3

What you are seeing is the transitions from one constellation point to another. In order to reduce the signal bandwidth, the baseband signal is low-pass filtered. This causes the transitions to not be instantaneous (i.e. the I and Q are not square waves), so they take some time. You are simply seeing those transitions. The low-pass filtering also causes ...


2

Ok, question solved. I found out that indeed L1 pre/post constellations are not rotated. The constellation above comprises: L1 pre (BPSK) L1 post (64-QAM) Data PLP from P2 symbol (rotated 16-QAM) Also P2 cells are interleaved (ETSI EN 302 755 @ 8.5).


2

so let's act as if we had only one active subcarrier "at a time", just to ignore interference between the different subcarriers. The frequency offset in time domain means a time offset in frequency domain, and probably a fractional one at that. Since that offset is constant for each carrier, the resulting sample stream (i.e. the output of your DFT) is ...


2

In my experience from the space industry, both near-earth and deep-space communication system resolve phase ambiguity by means either unique word detection or different flavors of differential encoding. In unique word detection technique, an "Attached Sync Marker (ASM)"(a.k.a a unique word), a preamble of some sort (usually the hex pattern 0x1ACFFC1D) is ...


2

The correlator is just a filtering operation. So in the case of passband pulse-amplitude modulation (PAM), such as QPSK or QAM, the (noiseless) received analytic signal before demodulation and sampling can be written as $$r(t)=e^{j((\omega_c+\Delta\omega) t+\phi))}\sum_{k}A_kp(t-kT)\tag{1}$$ where $\Delta\omega$ is a frequency offset, $\phi$ is a phase ...


2

Despite the great answers by Dan and Olli - I am convinced this is indeed just plain ISI. In my case it's introduced by the droop of the analog filters after the DAC cutting into the passband of my signal. It's pretty clear from my tests that by extending the analog filter cut-off I can remove this effect. Thinking about how that works, the highest frequency ...


2

If there is capacitance between signal and ground, a resistor-capacitor (RC) filter may be formed. It can also be an intentionally added filter, like in original poster's answer. An RC low-pass filter has an exponentially decaying impulse response. Due to the exponential decay, if the interference from a first symbol to the second symbol is 10 %, then the ...


2

I don’t think this is necessarily additionally introduced ISI (beyond the ISI of the pulse shaping filter itself, which is zero when there is no timing offset), but may be the result of timing offset error. All the points within a small offset from the ideal sampling location appear to be selected, and the overall pattern of the full constellation showing ...


1

CFO is modeled like this $x(t) = s(t) e^{j2\pi f_{\text{CFO}}t} $. This is a phase rotation only and does not effect the magnitude. Convince yourself of it by breaking $e^{j2\pi f_{\text{CFO}}t}$ up as it magnitude ($= 1$) and its phase ($=2\pi f_{\text{CFO}}$). This is all happening in the time domain. If you plot the CFO-free signal and the CFO-corrupted ...


1

You would need to recover some timing information from the transmitted signal. A common way is to transmit some sort of preamble before the rest of the data in your frame that is transmitted. The preamble would contain known symbols, so it could be used for multiple purposes, including channel estimation and timing recovery (also known as synchronization). ...


1

The following depicts a QPSK Carrier Recovery loop that works on the baseband I and Q samples following proper decisions from a Timing Recovery Loop. For more information on this implementation see High modulation index PSK - carrier recovery And for more information on a Timing Recovery implementation that can precede this see: Symbol timing ...


1

In the frequency domain a signal with a carrier offset looks like the following- This is usually modeled as the desired baseband signal convolved (in the frequency domain) with a complex tone with frequency equal to the carrier offset. As you are probably already aware, convolution in the frequency domain is equivalent to multiplication in the time domain. ...


1

You're right that the phase plot doesn't match the signal plot. Note that it also shows more symbols than the signal plot. Those two plots do not refer to the same bit sequence, and the phase plot is only meant to illustrate the difference between QPSK and OQPSK (the original figure consists of two plots, one for QPSK and one for OQPSK).


1

You've got it – zero mean noise still has power. That power happens to be its variance. RMS of such signals thus happens to be the noise amplitude's standard deviation – and give a sensible number to assess link quality and error probabilities.


1

Because the decimate function does not try to keep power constant. Remember that decimation is not inherently an LTI operation; so, probably, the authors of that function simply did not care too much for keeping the amplitude of a passband signal constant. Note that a decimation step typically includes filtering to $\frac1{\text{ds_factor}}$ of the ...


1

If the signal breaks the Nyquist criterion we'll only get a part of the signal. This needs to be clarified: the frequencies above Nyquist will fold back into the Nyquist band, so it's not just that you lose part of the signal; it's that the signal is corrupted (usually) beyond hope of reconstruction. As an example, consider a 4 kHz real signal sampled at 4 ...


1

As others have mentioned, it sounds like you are talking about the null carriers in an OFDM system: These bins are used to mitigate leakage from adjacent RF bands 2. Perhaps your tutor meant ISI in the sense that if adjacent RF bands were to spill into your OFDM signal, you would have some interference with your symbols on the edge of your bandwidth, but ...


1

Even though have AGC etc in your system, it is not ensured that the received constellation points will lie exactly at the points that you would expect from the TX constellation. There can be residual offset, gain and phase rotation (at least). In the following , there's a simple method to estimate and remove offset and gain. Essentially, you calculate the ...


1

The function you are using is from the Communication Systems Design Toolbox. You might have it not installed/licensed on your PC. In case you just want to plot a constellation diagram, you can go with this code (Note, that it works for QAM constellations, not BPSK (since BPSK has only one domain)). values = -7:2:7; % Values in real and complex domain [X, ...


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