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I am using Python to generate a sine wave in order to cancel out part of a signal I'm analyzing. I would like the magnitude of this signal in the FFT to be equal to the magnitude of the signal that I want to get rid of. My questions are:

  • A. How does the amplitude of a signal translate to the amplitude of the resultant FFT?

  • B. Is there a way I can easily control the amplitude of an FFT using the amplitude of an input signal?

I can post my code if it helps.

This is the FFT of the signal I'm analyzing. I want the large amplitude at 1.4-ish(from the signal I generated) to be equal in magnitude to the large amplitude at 2.9-ish (the frequency I want to get rid of)

This is the FFT of the signal I'm analyzing. I want the large amplitude at 1.4-ish(from the signal I generated) to be equal in magnitude to the large amplitude at 2.9-ish (the frequency I want to get rid of)

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    $\begingroup$ You need to consider sampling period $Ts$ window length $L$ and DFT size $N$ for computing the associated scaling that applies to the continuous time Fourier transform of a given signal while its DFT is being computed and plotted. $\endgroup$ – Fat32 Jun 20 '17 at 15:25
  • $\begingroup$ Code is usually helpful if you want specific answers. $\endgroup$ – Stephen Rauch Jun 20 '17 at 15:51
  • $\begingroup$ Can you use a notch filter to get rid of the signal of dis-interest? $\endgroup$ – jeremy Jun 21 '17 at 21:00
  • $\begingroup$ This question was already answered here $\endgroup$ – Andrei Keino Oct 7 '17 at 20:11
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Consider a continuous-time signal $$x_a(t) = A \cos(\Omega_0 t + \theta) $$ where $\Omega_0$ is the continuous time frequency in radians per second and $\theta$ is the phase in radians.

In order to perform a digital spectrum analysis of the continuos time signal $x_a(t)$ first sample (without aliasing) it with a period of $T_s$ seconds per sample (or with a sampling frequency of $F_s = 1/ T_s$ samples per second) resulting in the discrete-time signal $$x[n] = A \cos( \omega_0 n + \theta)$$ where $\omega_0 = \Omega T_s$ is the discrete-time frequency is radians per sample.

Next consider the relationship between the CTFT $X_a(\Omega)$ of $x_a(t)$ and DTFT $X(e^{j\omega})$ of $x[n]$ through the continuous to discrete time conversion process stated as: $$X(e^{j\omega}) = \frac{1}{T_s} X_a(\frac{\omega}{T_s})$$ which shows the relation over the first period of $X(e^{j\omega})$

Computation of the DTFT $X(e^{j\omega})$ of the sequence $x[n]$ may require indefinetely large number of samples (for example infinite if $x_a(t)$ is the ideal sine wave). However that's not practical for not only that it requires an incomprehensible amount of processing but, more importantly, also for it's not the DTFT $X(e^{j\omega})$ which is represented within the digital computer but its's DFT samples $X[k]$ which is defined over a finite length of signal.

Therefore, in order to get a finite length truncation of the input signal $x[n]$ apply a window: $$v[n] = w[n]x[n]$$ where $w[n]$ is the window of length $L$.

The consequence of this windowing on the result of the computed spectrum is: $$V(e^{j\omega}) = \frac{1}{2\pi} X(e^{j\omega}) \star W(e^{j\omega}) $$

Computing the $N$-point DFT of the sequence $v[n]$ therefore provides the samples $V[k]$ of the result of the periodical convolution between the true spectrum $X(e^{j\omega})$ of $x[n]$ and the DTFT $W(e^{j\omega})$ of the window $w[n]$. Note that $N \geq L$.

For the specific example we have considered, $$x[n]=A\cos(\omega_0 n + \theta)$$ whose DTFT is $$X(e^{j\omega})= A\pi e^{j\theta} \delta(\omega-\omega_0) + A\pi e^{-j\theta} \delta(\omega+\omega_0)$$ a pair impulses weighted by $A\pi$.

And assuming a rectangular window of length $L$ for $w[n]$, its DTFT is: $$W(e^{j\omega})= \sum_{n=0}^{L-1} w[n] e^{-j\omega n} = e^{-j{\omega (L-1)/2}} \frac{\sin(\omega L / 2)}{\sin(\omega / 2)} $$ the result is $$V(e^{j\omega}) = 0.5 A e^{j\theta} W(e^{j(\omega - \omega_0)}) + 0.5 A e^{-j\theta} W(e^{j(\omega + \omega_0)})$$

Finally the DFT samples $V[k]$ are given by the following: $$V[k] = V(e^{j\frac{2\pi k}{N} }) = 0.5 A e^{j\theta} W(e^{j(\frac{2\pi k}{N} - \omega_0)}) + 0.5 A e^{-j\theta} W(e^{j(\frac{2\pi k}{N} + \omega_0)}) $$

One peak peak of these DFT samples would occur for $\omega = \omega_0$ for some $k$ if $\frac{2\pi k}{N} = \omega_0$ is satisfied for an integer $N$ and $k$. Assuming $T_s$ ,$L$,$N$,and $\Omega_0$ are selected such that the equality is satisfied for some integer $k_0$, then the magnitude of the peak of the observed spectrum is: $$ |V[k_0]| = | 0.5 A e^{j\theta} W(e^{j(0)}) + 0.5 A e^{-j\theta} W(e^{j(2\omega_0)}) | $$

Solving for $A$, the amplitude of the original analog signal whose spectrum analysis are after, yields: $$A = \frac{2 |V[k_0]|}{|e^{j\theta} W(e^{j(0)}) + e^{-j\theta} W(e^{j(2\omega_0)}) |} $$

An approximation to this can be obtained by assuming, for the rectangular window, $W(e^{j 2\omega_0}) \approx 0$ and noting that $W(e^{j0}) = L$ : $$A \approx \frac{2 |V[k_0]|}{L}$$

If a window other than rectangular is used, formulation should be adjusted accordingly, specifically $W(0)$ being the sum of the window samples, which is $L$ for the rectangular window.

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